Full duplex reconfigurable antenna self-interference cancellation systems

ABSTRACT

Embodiments of transceivers with one or more reconfigurable antennas are described. In one embodiment, a reconfigurable antenna transceiver includes a transmit chain, a receive chain, and a reconfigurable antenna having a plurality of reconfigurable modes. The transceiver may also include an antenna controller configured to set a mode of the reconfigurable antenna. According to other aspects, the transceiver may also include a signal processor configured to transmit a set of training symbols during a training interval. The antenna controller may be further configured to select a respective mode of the reconfigurable antenna for each training symbol in the set of training symbols. Additionally, the antenna controller may be configured to calculate a received Signal-of-Interest to Interferer Ratio (SIR) for each training symbol of the set of training symbols. In this context, a system utilizing a reconfigurable antenna may achieve significant rate improvement compared to half-duplex systems.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a continuation of U.S. Non-Provisional patentapplication Ser. No. 14/576,295, filed Dec. 19, 2014, and claims thebenefit of U.S. Provisional Application No. 62/002,517, filed May 23,2014, the entire contents of both of which are hereby incorporatedherein by reference.

This invention was made with government support under grant ECCS-0955157awarded by the National Science Foundation. The government has certainrights in the invention.

BACKGROUND

Many wireless systems operate in a half-duplex communications mode,where wireless devices are either transmitting or receiving, but notusing the same temporal and spectral resources. Full-duplexcommunications modes may double the efficiency of bidirectionalcommunications over the same temporal and spectral resources. Infull-duplex mode, a wireless device can transmit radio frequency (RF)signals at a carrier frequency using a transmit antenna whilesimultaneously receiving RF signals over the same carrier frequencythrough a collocated receive antenna. A wireless device can alsotransmit RF signals at a carrier frequency using an antenna whilesimultaneously receiving RF signals over the same carrier frequencythrough the same antenna using a circulator or similar device. Onelimitation impacting full-duplex communications within a transceiver,however, is managing any self-interference signals imposed on thereceive antenna by the transmit antenna.

BRIEF DESCRIPTION OF THE DRAWINGS

For a more complete understanding of the embodiments described hereinand the advantages thereof, reference is now made to the followingdescription, in conjunction with the accompanying figures brieflydescribed as follows:

FIG. 1 illustrates a two antenna full-duplex reconfigurable antennatransceiver according to one example embodiment described herein.

FIG. 2 illustrates a two antenna full-duplex reconfigurable receiveantenna transceiver according to another example embodiment describedherein.

FIG. 3 illustrates a two antenna full-duplex reconfigurable transmitantenna transceiver according to another one example embodimentdescribed herein.

FIG. 4 illustrates a block diagram of a multi-antenna, multi-chainfull-duplex reconfigurable antenna transceiver according to anotherexample embodiment described herein.

FIG. 5 illustrates an example structure of a frame for maximizing theSignal-of-Interest to Interferer Ratio (SIR) according to theembodiments described herein.

FIG. 6A illustrates a cross-sectional side profile view of an examplemulti-reconfigurable antenna (MRA) according to one embodiment describedherein.

FIG. 6B illustrates a top-down view of the example MRA in FIG. 6Aaccording to one embodiment described herein.

FIG. 7 illustrates simulated and measured reconfigurable antennaradiation patterns for four different modes of operation of the MRA 600in FIGS. 6A and 6B, showing agreement between simulated and measuredpatterns.

FIG. 8 illustrates a representative example full-duplex communicationssystem consisting of two nodes according to the embodiments describedherein.

FIG. 9 illustrates an example floor plan for experiments.

FIG. 10 illustrates a Cumulative Distribution Function (CDF) of passiveself-interference suppression for the MRA system in FIG. 8 as comparedto a omni-directional antenna system.

FIG. 11 illustrates a CDF of the signal-of-interest power loss for threedifferent experimental environments in addition to the average CDF forall environments.

FIG. 12 illustrates a CDF of optimum pattern indexes according to theembodiments described herein.

FIG. 13 illustrates the number of reconfigurable antenna patternscapable of achieving a certain amount of passive suppression in at leastone tested scenario.

FIG. 14 illustrates passive self-interference suppression andsignal-of-interest power loss CDFs for various subsets of MRA modes.

FIG. 15 illustrates an achieved average passive self-interferencesuppression at different re-training times for semi-static and dynamicenvironments.

FIG. 16 illustrates residual self-interference power before and afterdigital cancellation at different transmit power values.

FIG. 17 illustrates achievable rate and rate gain for the full-duplexand half-duplex systems at different transmit power values.

FIG. 18 illustrates a flow diagram for an antenna mode reconfigurationprocess performed by the reconfigurable receive antenna transceiver inFIG. 2 according to an example embodiment.

FIG. 19 illustrates an example schematic block diagram of a processingenvironment which may be relied upon, in part, in one or more of thetransceivers in FIGS. 1-4, according to various embodiments describedherein.

The drawings illustrate only example embodiments and are therefore notto be considered limiting of the scope described herein, as otherequally effective embodiments are within the scope and spirit of thisdisclosure. The elements and features shown in the drawings are notnecessarily drawn to scale, emphasis instead being placed upon clearlyillustrating the principles of the embodiments. Additionally, certaindimensions may be exaggerated to help visually convey certainprinciples. In the drawings, similar reference numerals between figuresdesignate like or corresponding, but not necessarily the same, elements.

DETAILED DESCRIPTION

Due to increases in wireless data traffic, one challenge for futurewireless systems is the utilization of the available spectrum to achievebetter data rates. Recently, full-duplex communications, wherebidirectional communications are carried out over the same temporal andspectral resources, have been introduced as a mechanism to potentiallydouble the spectral efficiency of wireless systems. One limitationimpacting full-duplex communications within a transceiver is managingany self-interference signals imposed on the receive antenna by thetransmit antenna. Full-duplex systems may achieve substantial rateimprovement over half-duplex systems when self-interference signals aremitigated.

Self-interference cancellation techniques may be generally divided intopassive suppression and active cancellation categories. In passivesuppression, a self-interference signal is suppressed in the propagationdomain before it is processed by receiver circuitry. In activecancellation techniques, a self-interference signal is mitigated bysubtracting a processed copy of a transmitted signal from a receivedsignal. Experimental and analytical results show that the mitigationcapability of active cancellation techniques is relatively limited,mainly due to transmitter and receiver radio circuit impairments. On theother hand, as compared to active cancellation techniques, passivesuppression techniques mitigate both the self-interference signal andthe transmitter noise associated with it. In addition, mitigating theself-interference signal before it is processed by the receivercircuitry decreases the effect of receiver noise and increases thedynamic range allocated for the desired signal, thus achieving betterperformance.

Passive self-interference suppression may be achieved through one or acombination of the following methods: (i) antenna separation, (ii)antenna isolation, (iii) antenna directionality, and (iv) antennapolarization. The applicability of each of these techniques depends onboth the application and the physical constraints of the system. Forexample, in mobile applications with small device dimensions, the levelof passive suppression achieved using antenna separation and isolationis relatively limited. In other systems where transmit and receiveantennas are not necessarily collocated (e.g., relay systems), antennaseparation and isolation may achieve relatively significant passivesuppression. In contrast to situations where relatively large antennaseparation is possible, the embodiments described herein focus on thedeployment of full-duplex communications where antenna separation isrelatively limited.

It is noted that the directional antennas used in passive suppressionsystems are generally single pattern directional antennas. The lack ofbeam steering capability in such antennas may affect signal-of-interestpower in certain scenarios (e.g., when the desired signal is coming fromthe opposite direction of the antenna). On the other hand, the antennare-configurability described herein may be relied upon to maximize aperformance metric, such as the received Signal-of-Interest toInterferer Ratio (SIR) metric, which represents a metric of goodperformance as described in further detail below.

In the context provided above, various full-duplex reconfigurableantenna systems are described herein. In one embodiment, a full-duplexreconfigurable antenna transceiver includes a transmit chain, a receivechain, and a reconfigurable antenna having a plurality of reconfigurablemodes. The transceiver may also include an antenna controller configuredto set a mode of the reconfigurable antenna. According to other aspects,the transceiver may also include a signal processor configured totransmit a set of training symbols during a training mode. The antennacontroller may be further configured to select a respective mode of thereconfigurable antenna for each training symbol in the set of trainingsymbols. Additionally, the antenna controller may be configured tocalculate a performance metric, such as a received Signal-of-Interest toInterferer Ratio (SIR) for each training symbol of the set of trainingsymbols, for example. In this context, a full-duplex system utilizing areconfigurable antenna may achieve a significant rate of improvementcompared to half-duplex systems.

According to other aspects described herein, by appropriatelycontrolling (e.g., reconfiguring) antenna properties of a reconfigurableantenna, a high degree of passive suppression can be achieved tofacilitate full-duplex communications. As used herein, a reconfigurableantenna is any antenna or antenna system having reconfigurableproperties which may be dynamically changed according to certain inputconfigurations. Among others, some example reconfigurable antennaproperties include: (i) antenna radiation pattern, (ii) antennapolarization, and (iii) antenna operating frequency. A reconfigurableantenna system may be embodied as a single antenna element or an arrayof antenna elements.

In one example embodiment described herein, a full-duplex systemutilizing a multi-reconfigurable antenna (MRA) with about 90% rateimprovement compared to half-duplex systems is described. According tovarious aspects, an MRA may be embodied as a dynamicmulti-reconfigurable antenna or antenna array structure that is capableof changing its properties according to certain input configurations. Anexperimental analysis, as described below, was conducted to characterizesystem performance using the MRA in typical indoor environments. Theanalysis was performed using a fabricated MRA having 4,096 configurableradiation patterns. Examples of the MRA-based passive self-interferencesuppression are detailed below along with an analysis of MRA trainingoverhead. In addition, a heuristic-based approach is proposed to reduceMRA training overhead. The results show that, at 1% training overhead, atotal of about 95 dB self-interference cancellation is achieved intypical indoor environments. The 95 dB self-interference cancellation isexperimentally shown to be sufficient for a 90% full-duplex rateimprovement compared to half-duplex systems. Further, the passiveself-interference suppression techniques described herein may be usedalone or combined with other active self-interference cancellationtechniques to achieve better performance.

According to other aspects of the embodiments, an MRA pattern selectionmechanism is described. The pattern selection mechanism is tailored toselect an optimum pattern among various available MRA patterns. Becausean MRA may have many radiation patterns, for example, one can select asuitable pattern that minimizes received self-interference power. Toseek the best overall system performance, a pattern selection mechanismthat maximizes the received SIR at the receiver input may be reliedupon. Using an MRA as a receive antenna in a full-duplex communicationssystem, the performance of MRA-based passive self-interferencesuppression is experimentally investigated, as described in furtherdetail below. The results presented below show that MRA-based passiveself-interference suppression can achieve an average of about 65 dB ofpassive self-interference suppression, with about a 45 dB SIR gaincompared to when an omni-directional antenna is used.

Additionally, since an MRA may be trained in order to select an optimalantenna mode, training time and overhead parameters are investigated. Inthis context, a heuristic-based approach is proposed to reduce thetraining overhead by selecting a small suboptimal set of patterns amongthe full set of MRA antenna modes. The results show that, using theproposed heuristic-based approach at 1% training overhead with asuboptimal set of 300 patterns, about 62 dBs of passive suppression maybe achieved with only about a 3 dB performance loss as compared to theoptimal case.

In another embodiment, a method of reconfiguring an antenna of atransceiver is described. The method of reconfiguring includestransmitting, with a transmit chain of the transceiver, a set oftraining symbols, selecting, with an antenna controller, a respectivemode of a reconfigurable antenna of the transceiver for each trainingsymbol in the set of training symbols, and calculating, with an antennacontroller, a received performance metric for each of the set oftraining symbols. The method may further include selecting a mode of thereconfigurable antenna for use during a data transmission interval basedon the performance metric of each of the set of training symbols. Inother aspects, selecting the mode of the reconfigurable antenna may befurther based upon a threshold performance criteria of the transceiver,as described in further detail below.

Finally, a complete full-duplex system with a combined MRA-based passivesuppression and conventional active self-interference cancellation ispresented. The overall system performance is evaluated in differentindoor environmental conditions. The results show that, at 1% trainingoverhead, a total of about 95 dB self-interference cancellation isachieved in typical indoor environments. The 95 dB self-interferencecancellation is experimentally shown to be sufficient for 90%full-duplex rate improvement compared to half-duplex systems at about 5dBm transmit power.

Turning to the figures, various aspects of the embodiments are describedin further detail.

FIG. 1 illustrates a two antenna full-duplex reconfigurable antennatransceiver 100 according to one example embodiment described herein. Inoperation, the transceiver 100 may transmit and receive data, at leastin part, in an overlapping time period and/or overlapping frequencychannel or range. The transceiver 100 in FIG. 1 includes digital signalprocessor 110, transmit chain 120, reconfigurable transmit (TX) antenna130, receive chain 140, reconfigurable receive (RX) antenna 150, andantenna controller 160. As also illustrated in FIG. 1, the transmitchain 120 includes a digital domain TX processor 122, adigital-to-analog converter (DAC) 124, and an analog domain TX processor126. The receive chain 140 includes an analog domain RX processor 142,an analog-to-digital converter (ADC) 144, and a digital domain RXprocessor 146.

The operation of the transceiver 100 may be configured, at least inpart, by the antenna controller 160. As illustrated in FIG. 1, theantenna controller 160 provides one or more control signals to one orboth of the reconfigurable TX antenna 130 and/or the reconfigurable RXantenna 150 (“the antennas 130 and 150”). In this way, the antennacontroller 160 may administer or control the electrical structures,characteristics, and/or properties of the antennas 130 and 150 overtime. In other words, according to aspects of the embodiments, theantenna controller 160 is configured to determine and define certainoperating parameters of the antennas 130 and 150 over time to optimizevarious performance metrics, as described herein. As input, the antennacontroller 160 may receive any one, two, three, or all four of thesignals 162, 164, 166, and 168, respectively, the output of thereconfigurable RX antenna 150, the output of the analog domain RXprocessor 142, the output of the ADC 144, and the output of the digitaldomain RX processor 146, as illustrated in FIG. 1.

The digital signal processor 110 may be embodied as any suitableprocessor for digital signals and is configured to modulate transmitsymbols to transmit data and demodulate receive symbols to receive data.Generally, the digital signal processor 110 may be configured tomodulate and demodulate data for wireless communications using anysuitable digital modulation technique, such as amplitude shift keying(ASK), frequency shift keying (FSK), minimum shift keying (MSK), phaseshift keying (PSK), quadrature amplitude modulation (QAM), etc., with orwithout the use of error coding and correction (e.g., cyclic coding,block coding, adaptive coding etc.), multiplexing (e.g., orthogonalfrequency-division multiplexing, etc.), and/or spread spectrumtechniques. In this context, the digital signal processor 110 may beembodied as a general- or specific-purpose processor optimized thoughhardware, software, or a combination of hardware and software fordigital signal processing. The digital signal processor 110 may includememory to store and execute programs (e.g., signal processingalgorithms, signal filtering algorithms, etc.) and memory to store data(e.g., constellation space, symbol, bit, etc. data). Additional detailsregarding the structure and function of the digital signal processor 110are described in further detail below with reference to FIG. 20.

The digital domain TX processor 122 may be embodied as any suitablesignal processor for baseband data processing. In this sense, thedigital domain TX processor 122 may include one or more baseband digitalfilters, interpolators, decimators, scalers, etc. In this context, amongother functions, the digital domain TX processor 122 may be configuredto filter, rate-adapt, and/or scale digital signals received from thedigital signal processor 110 and, thus, prepare them fordigital-to-analog conversion and transmission over the antenna 130. TheDAC 124 may be embodied as any suitable digital-to-analog converterconfigured to convert a digital signal to an analog signal. The analogdomain TX processor 124 may be embodied as any suitable physical layerfront-end circuitry for wireless data transmission. In this sense, theanalog domain TX processor 124 may include one or more filters,frequency-upconverters, amplifiers, etc. In this context, among otherfunctions, the analog domain TX processor 124 is configured to amplify,frequency-upconvert, and transmit digitally-modulated data signals overthe antenna 130.

The analog domain RX processor 142 may be embodied as any suitablephysical layer front-end circuitry for wireless data reception. In thissense, the analog domain RX processor 142 may include one or morefilters, amplifiers, etc. In this context, among other functions, theanalog domain RX processor 142 is configured to amplifydigitally-modulated data signals received over the antenna 150. The ADC144 may be embodied as any suitable analog-to-digital converterconfigured to convert an analog signal to a digital signal. The digitaldomain RX processor 146 may be embodied as any suitable signal processorfor front-end data reception. In this sense, the digital domain RXprocessor 146 may include one or more digital filters, interpolators,decimators, scalers, etc. In this context, among other functions, thedigital domain RX processor 146 may be configured to filter, rate-adapt,and/or scale digital signals received from the ADC 144 and, thus,prepare them for further processing by the digital signal processor 110.

It should be appreciated that the transceiver 100 in FIG. 1 is providedby way of example only. That is, the passive self-interferencesuppression techniques described herein may be used in connection withother receiver, transmitter, and transceiver systems and architectures.As one example, FIG. 2 illustrates a two antenna full-duplexreconfigurable receive antenna transceiver 200 according to anotherexample embodiment described herein. As compared to the transceiver 100in FIG. 1, the transceiver 200 in FIG. 2 includes a non-reconfigurableTX antenna 132 in place of the reconfigurable transmit TX antenna 130.In the transceiver 200, the antenna controller 160 provides one or morecontrol signals to the reconfigurable RX antenna 150. In this way, theantenna controller 160 may administer or control the electricalstructure, characteristics, or properties of the antenna 150 over time.

As another example, FIG. 3 illustrates a two antenna full-duplexreconfigurable transmit antenna transceiver 300 according to anotherexample embodiment described herein. As compared to the transceiver 100in FIG. 1, the transceiver 300 in FIG. 3 includes a non-reconfigurableRX antenna 152 in place of the reconfigurable transmit RX antenna 150.In the transceiver 300, the antenna controller 160 provides one or morecontrol signals to the reconfigurable TX antenna 130. In this way, theantenna controller 160 may administer or control the electricalstructure, characteristics, or properties of the antenna 130 over time.

In other embodiments, any of the architectures illustrated in FIGS. 1-3may be extended to multi-antenna, multi-chain full-duplex systems with Mtransmit and N receive chains, for example, connected to one or morereconfigurable antennas, as illustrated in FIG. 4. FIG. 4 illustrates amulti-antenna, multi-chain full-duplex reconfigurable antennatransceiver 400. As compared to the transceiver 100 in FIG. 1, thetransceiver 400 in FIG. 4 includes two or more transmit chains (e.g.,120A, 120B, . . . ) similar to the transmit chain 120 in FIG. 1 and twoor more receive chains (e.g., 140A, 140B, . . . ) similar to the receivechain 140 in FIG. 1.

Generally, the reconfigurable antenna transceivers 100, 200, 300, 400,etc. described herein have different modes of operation. Each mode ofoperation is defined in part by certain electrical structures,characteristics, and/or properties (e.g., operating frequency,polarization, and radiation pattern, etc.) of one or both of theantennas 130 and 150. Referring among FIGS. 1-4, the antenna controller160 is configured to select and control the antenna mode of one or bothof the antennas 130 and 150. Based on the selected antenna mode andenvironmental conditions (e.g., frequency-selective fading, multipathinterference, etc.), which may vary over time, each mode achievescertain performance criteria that vary from others. In this context, theantenna controller 160 is configured to select an antenna mode thatsatisfies certain threshold performance requirements or criteria, forexample.

For further context, assume a full-duplex system with a reconfigurablereceive antenna and an omnidirectional transmit antenna. In this system,the self-interference signal and signal-of-interest may arrive at thereceive antenna from different directions of arrival depending upon theenvironmental conditions. Because the reconfigurable antennas describedherein are capable of changing their radiation pattern, for example, oneway to reduce the received self-interference power is to select aradiation pattern having a relatively lower antenna gain in thedirection of the self-interference signal. Thus, the antenna controller160 may be configured to select a radiation pattern having a relativelylower antenna gain in the direction of the self-interference signal,based on any one, two, three, or all four of the signals 162, 164, 166,and 168 in FIGS. 1-4. Generally, the antenna controller 160 may beconfigured to select an antenna mode that optimizes certain performancemetrics.

Since one goal of the transceivers described herein is to minimize anyreceived self-interference signal, the antenna controller 160 maymonitor received self-interference power, for example, as one metric forimprovement in performance. In this case, the antenna controller 160 mayselect an antenna mode that minimizes received self-interference power.However, because the antenna mode affects both the receivedself-interference and signal-of-interest simultaneously, minimizing thereceived self-interference power is not the only performance metric orfactor in achieving optimal performance. For instance, certain antennamodes may suppress the self-interference signal but also significantlyreduce the received signal-of-interest power. Thus, in full-duplexsystems, it should be appreciated that a more suitable performancemetric may be measured by way of received Signal-of-Interest toInterferer Ratio (SIR). SIR is defined as the ratio between the receivedsignal-of-interest power and the received self-interference power. Thus,the antenna controller 160 may select an antenna mode that maximizes thereceived SIR. According to one embodiment described herein, the antennacontroller calculates the SIR for each of L available antenna modes, andthen selects an antenna mode that maximizes the received SIR.

To assist the antenna controller 160 with the selection of an antennamode, a transmission frame may be divided into two main intervals: (i) atraining interval and (ii) a data transmission interval. During thetraining interval, the transceiver 100 (or one of the transceivers 200,300, or 400) transmits a training frame including a number of trainingsymbols equal to the number of antenna modes of one or a combination ofthe antennas 130 and 150. The training frame is processed by thetransmit chain 120 and transmitted by the antenna 130. Overlapping intime with the transmission of the training frame, the transceiver 100receives the training frame, as a self-interference signal, over theantenna 150, and the receive chain 140 processes it.

While the training frame is being transmitted and received from theantenna 130, the antenna controller 160 varies the mode of the antenna130, the antenna 150, or both the antennas 130 and 150 with eachtraining symbol. In this way, each transmitted training symbolcorresponds to a respective transmit and/or receive antenna mode. Itshould be appreciated that, for the transceiver 100 in FIG. 1, thecombined number of antenna modes may be determined as the number oftransmit modes of the antenna 130 multiplied by the number of receivemodes of the antenna 150. On the other hand, for the transceiver 200 inFIG. 2, the number of antenna modes is reduced to the number of receivemodes of the antenna 150 because the antenna 132 is not an MRA antennaand has only one transmit mode. Also, for the transceiver 300 in FIG. 3,the number of antenna modes is reduced to the number of transmit modesof the antenna 130 because the antenna 152 is not an MRA antenna and hasonly one receive mode.

FIG. 5 illustrates an example structure of a frame 500A for maximizingthe SIR according to the embodiments described herein. The frame 500A inFIG. 5 is provided by way of example only, as other frame structures maybe used to calculate SIR and are within the scope of the embodiments. InFIG. 5, the frame 500A includes a training interval 502A and a datatransmission interval 504A. During the training interval 502A, thetransceiver 100 transmits L training symbols (Symbol 1-Symbol L.) viathe antenna 130, where L is the number of antenna modes. That is, in oneembodiment, the transceiver 100 transmits L training symbols for eachmode of the antennas 130 and 150.

Each training symbol in the training interval 502A includes a gapinterval 512 (512A, 522A, 532A, etc.), a data interval 514 (514A, 524A,534A, etc.), and a null interval 516 (516A, 526A, 536A, etc.). The gapinterval 512 may be used to account for antenna switching time. Duringthe data interval 514, the transceiver 100 transmits a trainingsequence. During the null interval 516, the transceiver 100 is silent(i.e., refrains from any transmission).

The position of the data 514 and null intervals 516 within each trainingsymbol may be assigned such that, at any point in time, only onefull-duplex transceiver (e.g., 100, 200, 300, 400, etc.) is transmittingdata and all other transceivers are silent. For example, where a systemcomprises of two full-duplex nodes or transceivers (nodes A and B) incommunication with each other, the data 514 and null intervals 516 maybe alternated between the two nodes. In this context, frame 500B of asecond (node B) transceiver includes a training interval 502B having agap interval (512B, 522B, 532B, etc.), a null interval (516B, 526B,536B, etc.), and a data interval (514B, 524B, 534B, etc.). Comparing theframes 500A and 500B in FIG. 5, the positions of the data 514 and nullintervals 516 are alternated in time so that only one of the node A ornode B transceivers are transmitting at a time.

The training symbols 1-L of the transmission interval 502A are receivedby the transceiver 100 via the antenna 150. Further, in the case of twofull-duplex nodes or transceivers, combined training symbols 1-L of thetransmission intervals 502A and 502B are received by the transceiver 100via the antenna 150. Each received training symbol contains aself-interference portion and a signal-of-interest portion. Using theself-interference and signal-of-interest portions, the antennacontroller 160 processes each received training symbol and calculates areceived SIR for one or more of the antenna modes of the antennas 130and 150. The SIR for the antenna modes may be calculated using one ormore of the signals 162, 164, 166, and 168, for example, as references.

At the end of the training interval 502A, antenna controller 160 isconfigured to select an antenna mode that maximizes the SIR for useduring the data transmission interval 504. After the training interval502A, the data transmission interval 504A begins. During the datatransmission interval 504A, normal full-duplex communications occur. Theuse of one training symbol 1-L for each antenna mode involves arelatively exhaustive search approach. According to other embodimentsdescribed below, certain techniques may be relied upon to reduce thesearch space. For example, the set of possible antenna modes may bedivided into multiple groups to achieve coarse and fine mode searchingmethodologies. In this context, the antennal controller 160 may beconfigured to categorize search spaces to accelerate the selection of asuitable antenna mode.

FIG. 6A illustrates a cross-sectional side profile view of an exampleMRA 600 according to one embodiment described herein. Either of theantennas 130 and/or 150 in FIG. 1 may be embodied as the MRA 600. Asdescribed in further detail below, the MRA 600 may be configured todynamically change its proprieties (e.g., radiation pattern,polarization, operating frequency, etc.) according to control signalsprovided by the antenna controller 160 (FIG. 1), for example. At theoutset, it is noted that the MRA 600 in FIG. 6 is provided by way ofexample, is not drawn to scale, and is not intended to be limiting ofthe scope of the embodiments. In other words, while the MRA 600illustrated in FIGS. 6A (and 6B) is described below as having a certainstructure, the embodiments described herein may rely upon reconfigurableantennas which vary in size, shape, materials, modes of operation, etc.,as compared to the MRA 600.

The working mechanism of the MRA 600, which is embodied as a drivenantenna and multiple parasitic pixel elements, can be described by thetheory of reactively controlled directive arrays. The direction of themain beam of the MRA 600 may be directed by proper reactive loading ofparasitic pixel elements of the MRA 600. In the MRA 600, proper reactiveloading corresponds to a specific geometry of the parasitic pixelelements, which is obtained by switching PIN diode switches betweencertain pairs of adjacent pixels ON or OFF. Switching the PIN diodeswitches ON or OFF, as described in further detail below, provides 4,096different modes of operation for the MRA 600, each with a uniqueradiation pattern. As part of an empirical analysis of the MRA 600, FIG.7 illustrates simulated and measured reconfigurable antenna radiationpatterns for four different modes of operation of the MRA 600 in FIGS.6A and 6B, showing agreement between simulated and measured patterns.

The MRA 600 employs an aperture-coupled feed mechanism for radiofrequency (RF) feeding. The MRA 600 includes a driven patch antenna 602and a driven patch 604. From the top-down view (i.e., as in FIG. 6B),the driven patch 604 may be relatively square in shape and have lengthand width dimensions of about 19.3×19.3 mm², for example, although anysuitable dimensions are within the scope of the embodiments. In otherembodiments, the driven patch 604 may be formed in other sizes andshapes, such as circular, polygon, irregular, microstrip, etc. Also, thedriven patch 604 may vary in thickness among embodiments for the desiredoperating characteristics.

The driven patch antenna 602 comprises a patch array of electricallyconfigurable pixels, as further described below. Individual pixels(602A, 602B, 602C, etc.) of the driven patch antenna 602 may beelectrically configured (i.e., coupled) in combination with each otherto dynamically vary the properties of the MRA 600. As described infurther detail below with reference to FIG. 6B, the driven patch antenna602 may include 4, 9, 16, or any other number of individual pixels, anynumber of which may be electrically configured in combination with eachother to dynamically vary the properties of the MRA 600. In thiscontext, one or more pixel control lines 601, which are electricallycoupled to the driven patch antenna 602 via plated through-holes, forexample, may be relied upon to electrically configure the individualpixels of the driven patch antenna 602 (i.e., see control nodes A-L inFIG. 6B). From the top-down view (i.e., as in FIG. 6B), each of theindividual pixels 602A, 602B, 602C, etc. of the patch antenna 602 may berelatively square in shape and have length and width dimensions of about15×15 mm², for example, although any suitable dimensions are within thescope of the embodiments. In other embodiments, the individual pixels602A, 602B, 602C, etc. may be formed in other sizes and shapes, such ascircular, polygon, irregular, microstrip, etc. Also, the individualpixels 602A, 602B, 602C, etc. may vary in thickness among embodimentsfor the desired operating characteristics.

In one embodiment, the driven patch 604 is designed to operate in thefrequency band of 2.4-2.5 GHz and is fed by a 50Ω microstrip line 606through an aperture (21.4×1.4 mm²) etched through the center of a commonground plane 608. As illustrated in FIG. 6, a feed layer 610, a patchlayer 612, and a pixel surface layer 614 are interposed among the drivenpatch antenna 602, the driven patch 604, and the microstrip line 606. Inone embodiment, the feed layer 610, the patch layer 610, and the pixelsurface layer 612 are formed using the ROGERS CORPORATION 4003C®substrate laminate (εr=3.55, tan δ=0.0027), although other suitabletypes of substrates may be relied upon among embodiments. Further, inthe MRA 600, a gap 616 is formed between the patch layer 610 and thepixel surface layer 612, and the gap 616 is also formed using theRO4003C® laminate. The driven patch antenna 602, driven patch 604, andcommon ground plane 608 may each be formed from metal, such as copper,at any suitable thickness for the desired operating characteristics.

The feed layer 610 is 0.508 mm thick (Xf=0.508 mm), the patch layer 612is 3.048 mm thick (Xp=3.048 mm), the pixel surface layer 614 is 1.524 mmthick (Xs=1.524 mm), and the gap 616 is 7.62 mm thick (Xg=7.62 mm),although any suitable thicknesses of the layers 610, 612, and 614 andthe gap 616 are within the scope of the embodiments. The layers 610,612, and 614 and the gap 616 may be approximately 90×90 mm² in width andlength, as further described below with reference to FIG. 6B.

FIG. 6B illustrates a top-down view of the example MRA 600 in FIG. 6Aaccording to one embodiment described herein. The MRA 600 may be formedto approximately 90×90 mm² in width and length, although other sizes arewithin the scope of the embodiments. As illustrated, the driven patchantenna 602 of the MRA 600 includes individual pixels 602A-I, each sizedto approximately 15×15 mm², arranged in a 9-pixel 3×3 array. As notedabove, in other embodiments, the MRA 600 may be formed using fewer orgreater pixels in other grid arrangements.

Electrically-actuated electrical couplings are provided between certainpairs of individual pixels 602A-I of the driven patch antenna 602. Asillustrated in FIG. 6A, electrically-actuated electrical couplings areprovided between pixels 602D and 602E, 602E and 602F, 602A and 602B,602B and 602C, 602G and 602H, 602H and 602I, 602D and 602A, 602A and602G, 602E and 602B, 602B and 602H, 602F and 602C, and 602C and 602I. Inother embodiments, electrically-actuated couplings may be includedbetween other pairs of the individual pixels 602A-I, such as between thepixels 602D and 602B or 602D and 602I, for example.

As illustrated in FIG. 6B, each electrically-actuated electricalcoupling includes a direct current (DC) grounding inductor, a PIN diode,a DC block capacitor, and an RF choke inductor. For eachelectrically-actuated electrical coupling, one of the control nodes A-L,fed by one of the pixel control lines 601 (FIG. 6A), is relied upon tobias the PIN diode to an ON/OFF state. More particularly, as provided inFIG. 6B between the pixels 602D and 602E, for example, the DC groundinginductor 630 is coupled between the pixels 602D and 602E, a seriescombination of the PIN diode 632 and the DC block capacitor 634 iscoupled between the pixels 602D and 602E, and the RF choke inductor 636is electrically coupled between the control node A and the electricalconnection between the series combination of the PIN diode 632 and theDC block capacitor 634.

Thus, certain pairs of the individual pixels 602A-I are electricallycoupled (e.g., connected or disconnected) together by switching the PINdiode switches in the electrically-actuated electrical couplings ON orOFF using control voltages applied to the control nodes A-L by way ofthe pixel control lines 601. In this context, the antenna controller 160(FIG. 1) may provide control signals over the pixel control lines 601 toset the operating mode of the MRA 600. This control by the antennacontroller 160 may be relied upon to change the geometry of theparasitic surface of the MRA 600, which, in turn, changes the currentdistribution and RF characteristic of the MRA 600.

In one embodiment, the self-resonant frequency (SRF) of the RF chokeinductors (e.g., ref. 636) may be chosen at about 2.5 GHz, making themhigh impedance in the industrial, scientific, and medical (ISM) band, tominimize the current and effect on the bias lines to antennaperformance. The DC grounding inductors (e.g., ref. 630) provide groundsfor DC biasing purposes. The SRF of these DC grounding inductors may bechosen at about the same as the RF choke inductors to maintain high RFimpedance between pixels. The DC block capacitors (e.g., ref. 634) maybe relied upon to properly bias the PIN diode switches (e.g., ref. 632)as shown in FIG. 6B. The SRF of DC block capacitors may be chosen atabout 2.5 GHz to provide low RF impedance in the ISM band. In this way,the effect of DC block capacitors on RF performance is minimized.

Turning to an experimental analysis of an example transceiver (e.g., thetransceiver 200 in FIG. 2) using the MRA 600 in FIGS. 6A and 6B, theperformance of the example transceiver was characterized in a typicalindoor environment using the MRA 600 with 4,096 dynamically configurableradiation patterns. Due to the dependence of full-duplex systemperformance on the surrounding environment, such an experimentalanalysis is important for performance characterization.

FIG. 8 illustrates a representative example full-duplex communicationssystem 800 according to the embodiments described herein. The system 800includes transceiver node A 810 and transceiver node B 820. As transmitantennas, the transceiver nodes 810 and 820 rely upon dipole antennas.As receive antennas, the transceiver node A 810 relies upon the MRA 600in FIGS. 6A and 6B, and the transceiver node B 820 relies upon a dipoleantenna. For both the transceiver nodes 810 and 820, the UniversalSoftware Radio Peripheral (USRP) software defined radio (SDR) platformwas relied upon. Each USRP contains a Radio Frequency (RF) transceiverand a Field Programmable Gate Array (FPGA). The USRPs were connected toa host computer through a Gigabit Ethernet connection. Baseband signalprocessing was performed on the host computer. The baseband signals werestreamed to and from the USRPs at a rate of 25M sample/sec. Allexperiments were performed in the ISM band at 2.5 Ghz carrier frequencywith a 10 Mhz signal bandwidth. All USRPs were synchronized to onereference clock. The antenna mode for the MRA 600 was achieved through a12-line digital control cable. The timing of all USRPs and the FPGA thatdrives the MRA antenna switches were aligned with a one reference pulseper second (PPS) signal. For comparison purposes, another full-duplexsystem having only omni-directional transmit and receive antennas wasalso used.

Two different frameworks, including a passive suppressioncharacterization framework and a complete system framework, were usedfor characterization of the performance of the system 800. In thepassive suppression characterization framework, the system 800 was usedto characterize the achieved passive self-interference suppression foreach MRA radiation pattern of the MRA 600 at different environmentalconditions. For measurement purposes, in this framework, received SIR isused as a performance metric. The frame structure used forcharacterization of the passive suppression is described above withreference to FIG. 5.

In the frame structure, each transmission frame consists of L trainingsymbols, where L is the number of antenna patterns or modes of the MRA600 to be characterized. Each training symbol contains three intervalsincluding a gap interval, a data interval, and a null interval. The MRAradiation pattern of the MRA 600 was changed at the edge of eachtraining symbol, and the gap interval was used to account for MRAradiation pattern switching time. At the receiver side of thetransceiver node A 810, the transmitted training symbols are combinedand received by the MRA 600. In the combined training symbols, eachsegment contains a self-interference portion and a signal-of-interestportion. The received signal strength is calculated for each portion toobtain an estimate for the received self-interference andsignal-of-interest power.

In the complete system framework, overall system performance ischaracterized when MRA-based passive self-interference suppression iscombined with conventional digital cancellation techniques. In thisframework, two different performance metrics are used, including overallself-interference cancellation and the achievable full duplex rate. Thetransmission frame structure in the complete system framework consistsof the MRA training interval 502A and the data transmission interval504A (FIG. 5). During the MRA training interval 502A, MRA patterns aretrained and an optimum pattern is selected. During the data transmissioninterval 504A, full-duplex data transmission takes place between the twonodes 810 and 820. The data transmission interval 504A consists ofseveral data frames that have a frame structure the same as or similarto that in IEEE 802.11n systems. Each data frame consists of severalOrthogonal Frequency Division Multiplexing (OFDM) symbols with 64subcarriers in each symbol. At the beginning of each data frame,training symbols are transmitted for channel estimation purposes. Afterchannel estimation, digital self-interference cancellation is performedto mitigate the residual self-interference signal.

Since the optimum pattern selection process involves training, trainingtime and training overhead design parameters are investigated. Accordingto the structure of the training interval 502A, the training time andoverhead are a function of the number of MRA patterns that have to betrained and the length of each training symbol in the training interval502A. The length or duration of each training symbol is a function ofthe lengths of the gap 512 and data intervals 514 (FIG. 5). The lengthof the gap interval 512 is directly proportional to the MRA switchingtime, which is function of the MRA switching circuitry of the MRA 600.In the current embodiment, MRA switching time is about 0.5 us. Thelength of the data interval 514 depends on how the received signalstrength is calculated. For example, if the received signal strength iscalculated in the digital domain, the ADC sampling rate and allowabletiming offset will determine the minimum data interval length. Based onexperiments, approximately 30 time-domain samples are enough to obtain agood estimate for received signal strength. Therefore, using a 40 MHzADC sampling rate, the required minimum segment duration is 2 us (0.5 usfor antenna switching, and 1.5 us for data and null intervals persegment). This time could be reduced to 1.25 us if the ADC sampling rateis doubled to 80 Mhz, which is a practical sampling rate in currentwireless systems.

FIG. 9 illustrates an example floor plan 900 for experiments. The floorplan 900 presents a typical laboratory environment with measurementworkstations, tables, metallic surfaces, etc. The outer walls of thebuilding are made of concrete or glass with steel pillars, while theinner walls are made of drywall with steel pillars. To enrich theexperimental analysis, the two communicating nodes are placed atdifferent positions inside and outside the laboratory to create avariety of Line Of Sight (LOS) and non-LOS environments. In addition,various MRA orientations were tested so that the two communicating nodeswere facing each other, opposite to each other, or side to side. Toemulate typical conditions, the experiments were performed in bothsemi-static and dynamic environments. In a semi-static environment, thearea was static with no moving personnel in the near area. In dynamicenvironments, normal laboratory activities were maintained with movingpersonnel during the experiment time.

Below, the performance of the full-duplex communications system 800(“the MRA system 800”), which relies upon MRA-based passive suppression,is described in further detail with reference to various charts. Theperformance is compared to a conventional omni-directional antenna basedpassive suppression system (“the conventional omni-directional system”).Additionally, a heuristic-based approach to reduce the overall MRAtraining time is described. The performance of the heuristic-basedapproach is compared to the optimal case where all MRA patterns aretrained. Finally, the MRA training overhead and training periodicity aredescribed. The passive suppression framework is used to characterize theachieved MRA-based passive self-interference suppression, andperformance is evaluated at different transmit power values ranging fromabout −10 dBm to 10 dBm. Each run lasts for several seconds. In eachrun, all 4,096 MRA patterns of the MRA 600 (FIG. 6) are trained, and thepattern that maximizes the SIR is selected.

FIG. 10 illustrates a Cumulative Distribution Function (CDF) of thepassive self-interference suppression for the MRA system 800 in FIG. 8as compared to an omni-directional antenna system. The passivesuppression is defined as the ratio between the transmit power and thereceived self-interference power at the antenna output. The CDF iscalculated over time for all different runs and transmit power values.The results show that the MRA system 800 achieves an average of about 65dB passive suppression, with about 45 dB passive suppression gaincompared to the omni-directional antenna system.

Since the selected MRA pattern affects the received signal-of-interestpower, the achieved passive suppression amount is not sufficient tocharacterize the overall system performance. Instead, the effect of theMRA on the received signal-of-interest power should be also considered.The received signal-of-interest power is affected by both the MRApattern and the distance between the two communicating nodes. Thus, toeliminate the distance factor and focus only on the MRA effect, thesignal-of-interest power loss is used as a performance metric instead ofthe absolute value of the received signal-of-interest power. Thesignal-of-interest power loss is defined as the receivedsignal-of-interest power ratio between the MRA case and theomni-directional antenna case for the same experimental environment.

FIG. 11 illustrates a CDF of the signal-of-interest power loss for threedifferent experimental environments in addition to the average CDF forall environments. In the opposite orientation environment, the back sideof the MRA at one node is facing the other node. The face-to-faceorientation is contrary to the opposite orientation. In the side-to-sideorientation, the side of the MRA at one node is facing the other node.The main difference between the opposite orientation and theface-to-face orientation is that, in the opposite orientation, the MRAis receiving most of the signal-of-interest power through its back loopswhich generally have small antenna gain. However, in the face-to-faceorientation, most of the power is received through the main loops of theMRA which generally have high gain due to antenna directivity. Thus, itis expected to have signal-of-interest power loss in the oppositeorientations, while in the face-to-face orientation, the MRA is supposedto achieve signal-of-interest power gain. As shown in FIG. 11, anaverage of about 5 dB loss in the signal-of-interest power is expectedin the opposite orientation environments, with an averagesignal-of-interest power gain of about 4 dB in face-to-face orientationsand about 1 dB in side-to-side orientations, respectively. Over allorientations, an average signal-of-interest power loss of about 1 dB isexpected when the MRA is used.

While using an MRA antenna may lead to significant gains in passivesuppression, the investment in training time required to the optimalmode(s) of the MRA may be relatively large. In this context, accordingto aspects of the embodiments, a heuristic-based approach is relied uponto reduce the training time overhead. To address this issue, thedistribution of the optimal MRA pattern over time and for differentenvironmental conditions was calculated. FIG. 12 illustrates a CDF ofoptimum pattern indexes according to the embodiments described herein.The calculated distribution may be used to check if the optimal patternindex is localized or spans the whole range from 1 to 4,096. The resultsin FIG. 12 show that the optimum pattern index spans the whole range,but it is not uniformly distributed. The results show that there aresome patterns that have low or even zero probability to be among theoptimum patterns, while other patterns have high probability to be amongthe optimum ones.

While one viable choice may be to exclude patterns with low probabilityof being optimal, it is important to take into account the degree ofsub-optimality. For a pattern to have a low (or zero) probability ofbeing optimum does not necessary mean that the pattern achieves poorperformance. For instance, among those low probability patterns thereare two categories: i) patterns that achieve good performance that areslightly less than the performance of the optimal pattern, and ii)patterns with poor performance that are significantly less than that ofthe optimal pattern. Although they have significant differences inperformance, the probability criterion does not differentiate betweenthe two categories, because they are both considered non-optimal.Accordingly, a better selection criterion should involve theself-interference suppression performance for each pattern and not onlythe probability of being among the optimum patterns.

For further clarification, consider that in full-duplex systems, theself-interference signal arrives at the receive antenna in two maincomponents: the line of sight (LOS) component through the direct linkbetween the transmit and receive antennas and the non-LOS component dueto the reflections. Due to the close proximity of the transmit andreceive antennas, the LOS component is much higher than the non-LOScomponent. Therefore, any MRA pattern with high gain in the LOSdirection will most likely achieve poor performance. As such, thispattern may be avoided. The optimal patterns are the patterns that arecapable of suppressing not only the LOS component but also part of thenon-LOS component.

Accordingly, based on the achieved self-interference suppression foreach MRA pattern, a heuristic-based approach was developed to select asuboptimal set of patterns that are expected to achieve the bestperformance. First, a system was run in 16 different environments thatincludes a variety of LOS, non-LOS, semi-static, and dynamic scenarios,each with 4 different orientations (opposite, face-to-face, and twoside-to-side orientations). In each run, the achieved passiveself-interference suppression for each one of the MRA modes wascalculated. A certain threshold X is set that represents a desiredpassive self-interference suppression amount. Then, the patterns thatachieve passive suppression>X at any time in any environment areselected. In other words, the patterns that are capable of achievingpassive suppression>X at least once are selected. Thus, any pattern thatis not selected should have passive suppression less than X in alltested scenarios.

FIG. 13 illustrates the number of reconfigurable antenna patternscapable of achieving a certain amount of passive suppression in at leastone tested scenario. The results in FIG. 13 show the number of patternscapable of achieving passive suppression>X at least once for differentvalues of the threshold X. For instance, the results show that there are1000 and 300 patterns capable of achieving passive suppression >52 dBand 58 dB, respectively.

In order to test the accuracy of the proposed heuristic-based approach,two different suboptimal set of patterns were selected with passivesuppression thresholds of X=52 dB and 58 dB, respectively. The first setcontains 1000 patterns and the second set contains 300 patterns. Theperformance of the selected sets were characterized in more than 20different experimental environments different from the 16 environmentsused to select the suboptimal sets. FIG. 14 illustrates passiveself-interference suppression and signal-of-interest power loss CDFs forvarious subsets of MRA modes. The results show that the 300 pattern setachieves an average of about 62 dB passive self-interference suppressionwith about 3 dB loss compared to the optimal 4,096 pattern set, but atabout 14 times less training time. Also, at about 4 times less trainingtime, the 1000 pattern set achieves an average of about 64 dB passiveself-interference suppression. On the other hand, fromsignal-of-interest perspective, the results show that the 1000 and 300pattern sets achieve almost the same performance as the optimal 4,096pattern set. It is noted that the experimental environments in thisanalysis are different from the environments used to select thesuboptimal sets in the sense that the positions of the two communicatingnodes are changed and different orientations spanning the 360 degreesare used.

In this analysis, experiments are conducted in two main environments:semi-static and dynamic. FIG. 15 illustrates achieved average passiveself-interference suppression at different re-training times for thesemi-static and dynamic environments. The conclusions from these resultsare multifold. First, due to the slow channel variations in thesemi-static environment, the system performance is almost constant withrespect to the re-training time. In this type of environment, the MRAcould be trained once per second with no performance loss. Assuming thateach pattern requires 2 us training time, the training duration for the4,096, 1000, and 300 pattern sets are about 8 ms, 2 ms, and 0.6 msrespectively. If the MRA is trained once per second, the trainingoverhead for the 4,096, 1000, and 300 pattern sets will be 0.8%, 0.2%,and 0.06% respectively, which is a relatively negligible overheadcompared to the expected capacity gain achieved by full-duplex systems.

Second, in the dynamic environment, due to the relatively fast channelvariations, the system starts to lose performance with the increase ofthe re-training time. The results show that 2-3 dB passiveself-interference suppression loss is expected when the re-training timeincreases from 50 ms to 500 ms. However, for fair comparison of thedifferent pattern sets, the overall training overhead should beconsidered. Thus, rather than focusing on the re-training time, it isdesired to observe performance at a fixed training overhead. Forexample, if the training overhead is fixed at 1% with a 2 us patterntraining interval, the 4,096, 1000, and 300 pattern sets should becompared at re-training times of about 800 ms, 200 ms, and 60 ms,respectively. Comparing the performance of the different sets at theprevious re-training times, we note that all different sets achieveapproximately the same performance.

Another practical aspect that should be considered for re-training timeis the useful data frame length. Although the performance of the optimum4,096 pattern set is best, for reasonable training overhead, therequired re-training time for the 4,096 pattern set is higher. Forinstance, from the previous examples, the optimal 4,096 pattern set at1% training overhead, a re-training time of 800 ms is requiredregardless of the useful data length transmitted within the 800 ms. Inother words, to guarantee a 1% training overhead, a useful data framelength of about 800 ms should be transmitted between the two successiveMRA training intervals. Thus, in a multi-user networks, each user shouldbe assigned a continuous 800 ms interval for data transmission, which isrelatively large interval. On the other hand, the 300 pattern setrequires only 60 ms re-training time. Accordingly, from a practicalperspective, using smaller pattern sets alleviates the constraints onthe overall network performance.

Below, the overall performance of a full-duplex system utilizing an MRAantenna is described. For full system performance characterization,MRA-based passive suppression is combined with the conventional digitalself-interference cancellation techniques. In the full-duplex system,the received signal in the time and frequency domains can be written as:

y _(n) =h _(n) ^(I)*(x _(n) ^(I) +z _(n) ^(T))+h _(n) ^(S)*(x _(n) ^(S)+z _(n) ^(T))+z _(n) ^(R),   (1)

Y _(k) =H _(k) ^(I)(X _(k) ^(I) +Z _(k) ^(T))+H _(k) ^(S)(X _(k) ^(S) +Z_(k) ^(T))+Z _(k) ^(R),   (2)

where x^(I), x^(S) are the transmitted time domain self-interference andsignal-of-interest signals, h^(I), h^(S) are the self-interference andsignal-of-interest channels, z^(T) represents the transmitter noise,z^(R) represents the receiver noise, n is the time index, k is thesubcarrier index, * denotes convolution process, and uppercase lettersdenote the frequency-domain representation of the correspondingtime-domain signals. The digital cancellation is performed bysubtracting the term Ĥ_(k) ^(I)X_(k) ^(I) from the received signal in(2). Ĥ^(I) is an estimate for the self-interference channel, obtainedusing training sequences transmitted at the beginning of each dataframe.

The analysis below characterizes, the overall self-interferencecancellation achieved using MRA-based passive suppression followed bydigital cancellation. The complete system framework is used tocharacterize the overall self-interference cancellation performance. Atthe beginning, the MRA is trained and the optimum pattern is selected.Then, a sequence of data frames are transmitted from one node and theother node remains silent. In this case, the received data framecontains only the self-interference signal and the noise associated withit. The self-interference channel is estimated at the beginning of eachdata frame and the digital cancellation is performed. The totalself-interference suppression is calculated as the ratio between thetransmit power and the residual self-interference power after digitalcancellation.

FIG. 17 illustrates residual self-interference power before and afterdigital cancellation at different transmit power values. The resultsshow that, in addition to the about 63 dB passive suppression, digitalcancellation could achieve up to about 32 dB more self-interferencecancellation for a total of about 95 dB self-interference cancellation.At high transmit power values, the 32 dB gain is mainly limited by thetransmitter noise which cannot be eliminated using conventional digitalcancellation techniques. On the other hand, at low transmit powervalues, the achieved digital cancellation amount is limited by thereceiver noise floor. At lower transmit power levels, theself-interference signal is totally suppressed to below the receivernoise floor and the full-duplex systems is expected to achieve nearly100% rate gain compared to half-duplex systems.

One important performance metric in full-duplex systems is theachievable rate gain compared to half-duplex systems. In this analysis,the achievable rate of the proposed full-duplex system is characterizedin different experimental environments at different transmit powervalues. The performance is compared to the half-duplex systemperformance in the same environments. The achievable rate is calculatedas a function of the effective Signal to Noise Ratio (SNR) as R−log 2(1+SNR). One way to calculate the effective SNR in experimental analysisis by calculating the Error Vector Magnitude (EVM), defined as thedistance between the received symbols (after equalization and digitalcancellation) and the original transmitted symbols. Using an EVM to SNRconversion method, the SNR is calculated as SNR=1/(EVM)².

The average achievable rate for both full-duplex and half-duplex systemsis calculated as:

$\begin{matrix}{{R^{FD} = {\frac{1}{NMK}{\sum\limits_{n = 1}^{N}{\sum\limits_{m = 1}^{M}{\sum\limits_{k = 1}^{K}{\log_{2}\left\lbrack {1 + {SINR}_{n,m,k}} \right\rbrack}}}}}},} & (3) \\{{R^{HD} = {\frac{1}{NMK}{\sum\limits_{n = 1}^{N}{\sum\limits_{m = 1}^{M}{\sum\limits_{k = 1}^{K}{\frac{1}{2}{\log_{2}\left\lbrack {1 + {SNR}_{n,m,k}} \right\rbrack}}}}}}},} & (4)\end{matrix}$

where R^(FD), R^(HD) are the average achievable rate for full-duplex andhalf-duplex systems, SINR is the effective signal to interferer plusnoise ratio in full-duplex system, SNR is the effective signal to noiseratio in half-duplex system, N, Al, and K are the total number of dataframes, OFDM symbols per frame, and subcarriers per OFDM symbol,respectively. The factor of ½ in the half-duplex rate equation is due tothe fact that each half-duplex node is transmitting only half of thetime.

FIG. 17 illustrates achievable rate and rate gain for full-duplex andhalf-duplex systems at different transmit power values. The results showthat the proposed full-duplex system achieves about 80-90% rate gaincompared to the half-duplex system at about 5 dBm transmit power intypical indoor environments. The reason why the proposed full-duplexsystem could not achieve the 100% rate gain even at low transmits powervalues is due to the 1 dB signal-of-interest power loss shown in FIG.15. This signal-of-interest power loss makes the full-duplex SINR lessthan the half-duplex SNR by about 1 dB even if the self-interferencesignal is totally suppressed below the noise floor. On the other hand,the performance difference between the 1000 and 300 pattern sets is dueto the difference in the achieved self-interference cancellation amountas shown in FIG. 17.

Before turning to the antenna mode reconfiguration process flow diagramof FIG. 18, it is noted that the process may be practiced using analternative order of the steps illustrated in FIG. 18. That is, theprocess flow is provided as an example only, and the embodiments may bepracticed using process flows that differ from that illustrated.Additionally, it is noted that not all steps are required in everyembodiment. In other words, one or more of the steps may be omitted orreplaced, without departing from the spirit and scope of theembodiments. Further, steps may be performed in different orders, inparallel with one another, or omitted entirely, and/or certainadditional steps may be performed without departing from the scope andspirit of the embodiments. Finally, although the process 1800 in FIG. 18is generally described in connection with the transceiver 200 in FIG. 2,the process 1800 may be performed by other transceivers (e.g., thetransceivers 100, 300, 400, etc.).

FIG. 18 illustrates a flow diagram for an antenna mode reconfigurationprocess 1800 performed by the reconfigurable receive antenna transceiver200 in FIG. 2 according to an example embodiment. At reference numeral1802, the process 800 includes selecting, with the antenna controller160 (FIG. 2), a mode of the reconfigurable antenna 150 of thetransceiver 200. The reconfigurable antenna 150 may, in someembodiments, be similar to the MRA 600 in FIGS. 6A and 6B. In otherwords, at reference numeral 1802, the antenna controller 160 mayelectrically bias one or more of the PIN diodes in the MRA 600, asdescribed above with reference to FIGS. 6A and 6B, for example.

At reference numeral 1804, the process 1800 includes transmitting, withthe transmit chain 120 of the transceiver 200, a training symbol duringpart of a training interval. The training symbol may be similar to oneof the L training symbols described above with reference to FIG. 5. Atreference numeral 1806, the process 1800 includes calculating, with theantenna controller 160, a performance metric associated with thetraining symbol transmitted at reference numeral 1804. Here, it is notedthat a portion of the training symbol transmitted at reference numeral1804 may be received as self-interference over the MRA 600. Thus, atreference numeral 1806, a performance metric associated with thereceived self-interference may be calculated based on a thresholdperformance criteria of the transceiver 200. Any of the performancemetrics or criteria described herein, such as SIR, may be calculated bythe antenna controller 160 at reference numeral 1806, depending uponvarious design considerations.

At reference numeral 1808, the transceiver 200 determines whetheranother mode of the MRA 600 is available for consideration. Withreference to the MRA 600 in FIG. 6A or 6B, for example, a total of 4,096dynamically configurable radiation patterns or modes are available.Thus, the process 1800 may cycle through all 4,096 radiation patterns ofthe MRA 600, to determine an optimal pattern. Alternatively, asuboptimal set of patterns that are expected to achieve the bestperformance may be cycled, as described above, to reduce training time.In either case, if another mode of the MRA 600 is available forconsideration, the process proceeds back to reference numeral 1802 forthe selection of another mode of the MRA 600 and the transmission ofanother training symbol at reference numeral 1804.

If another mode of the MRA 600 is available for consideration, theprocess 1800 proceeds from reference numeral 1808 to 1810. At referencenumeral 1810, the process 1800 includes selecting, with the antennacontroller 160, a mode of the MRA 600 for use during a data transmissioninterval. The mode may be selected based on the performance metrics ofthe training symbols calculated at reference numeral 1806. Finally, atreference numeral 1820, the process 1800 includes beginning a datatransmission interval for the transceiver 200 using the antenna modeselected at reference numeral 1810.

FIG. 19 illustrates an example schematic block diagram of a processingenvironment 1900 which may be relied upon, in part, in one or more ofthe transceivers 100, 200, 300, or 400 in FIGS. 1-4, according tovarious embodiments described herein. For example, the processingenvironment 1900 may form part of the digital signal processor 110, thetransmit chain 120, and/or the receive chain 140 in one or more of thetransceivers 100, 200, 300, or 400 in FIGS. 1-4. The processingenvironment 1900 may be embodied, in part, using one or more elements ofa mixed general and/or specific purpose computer. The processingenvironment 1900 includes a processor 1910, a Random Access Memory (RAM)1920, a Read Only Memory (ROM) 1930, a memory device 1940, and an InputOutput (I/O) interface 1950. The elements of processing environment 1900are communicatively coupled via one or more local interfaces 1902. Theelements of the processing environment 1900 are not intended to belimiting in nature, as the architecture may omit elements or includeadditional or alternative elements.

In various embodiments, the processor 1910 may be embodied as one ormore circuits, general purpose processors, state machines, ASICs, or anycombination thereof. In certain aspects and embodiments, the processor1910 is configured to execute one or more software modules which may bestored, for example, on the memory device 1940. The software modules mayconfigure the processor 1910 to perform the tasks undertaken by one ormore of the transceivers 100, 200, 300, or 400 in FIGS. 1-4. In certainembodiments, the process 1800 described in connection with FIG. 18 maybe implemented or executed by the processor 1910 according toinstructions stored on the memory device 1940.

The RAM and ROM 1920 and 1930 may include or be embodied as any randomaccess and read only memory devices that store computer-readableinstructions to be executed by the processor 1910. The memory device1940 stores computer-readable instructions thereon that, when executedby the processor 1910, direct the processor 1910 to execute variousaspects of the embodiments described herein.

As a non-limiting example group, the memory device 1940 includes one ormore non-transitory memory devices, such as an optical disc, a magneticdisc, a semiconductor memory (i.e., a semiconductor, floating gate, orsimilar flash based memory), a magnetic tape memory, a removable memory,combinations thereof, or any other known non-transitory memory device ormeans for storing computer-readable instructions. The I/O interface 1950includes device input and output interfaces, such as keyboard, pointingdevice, display, communication, and/or other interfaces. The one or morelocal interfaces 1902 electrically and communicatively couples theprocessor 1910, the RAM 1920, the ROM 1930, the memory device 1940, andthe I/O interface 1950, so that data and instructions may becommunicated among them.

In certain aspects, the processor 1910 is configured to retrievecomputer-readable instructions and data stored on the memory device1940, the RAM 1920, the ROM 1930, and/or other storage means, and copythe computer-readable instructions to the RAM 1920 or the ROM 1930 forexecution, for example. The processor 1910 is further configured toexecute the computer-readable instructions to implement various aspectsand features of the embodiments described herein. For example, theprocessor 1910 may be adapted or configured to execute the process 1800described above in connection with FIG. 18. In embodiments where theprocessor 1910 includes a state machine or ASIC, the processor 1910 mayinclude internal memory and registers for maintenance of data beingprocessed.

A full-duplex system utilizing one or more MRAs is described herein. Thedescribed MRA is a reconfigurable antenna capable of dynamicallychanging its properties according to certain input configurations. Theperformance of the MRA system is experimentally investigated indifferent indoor environments. The results show that a total of about 95dB self-interference cancellation may be achieved by combining theMRA-based passive suppression technique with conventional digitalself-interference cancellation techniques. In addition, the full-duplexachievable rate is experimentally investigated in typical indoorenvironments showing that the proposed full-duplex system achieves up toabout 90% rate improvement compared to half-duplex systems in typicalindoor environments.

Although embodiments have been described herein in detail, thedescriptions are by way of example. The features of the embodimentsdescribed herein are representative and, in alternative embodiments,certain features and elements may be added or omitted. Additionally,modifications to aspects of the embodiments described herein may be madeby those skilled in the art without departing from the spirit and scopeof the present invention defined in the following claims, the scope ofwhich are to be accorded the broadest interpretation so as to encompassmodifications and equivalent structures.

1. A transceiver, comprising: a transmit chain configured to transmitover a first antenna; a receive chain configured to receive over asecond antenna, wherein at least one of the first antenna and the secondantenna comprises a reconfigurable antenna having plurality ofreconfigurable antenna modes; a digital signal processor configured totransmit a set of training symbols over the first antenna during atraining interval; and an antenna controller configured to: to select arespective mode of the reconfigurable antenna for individual trainingsymbols in the set of training symbols during the training interval; andset a mode of the reconfigurable antenna after the training interval. 2.The transceiver of claim 1, wherein the antenna controller is furtherconfigured to select the mode of the reconfigurable antenna based on athreshold performance metric of the transceiver during a datatransmission interval.
 3. The transceiver of claim 1, wherein a numberof the set of training symbols is equal to a number of the plurality ofreconfigurable antenna modes of the reconfigurable antenna.
 4. Thetransceiver of claim 1, wherein a number of the set of training symbolsis less than a number of the plurality of reconfigurable antenna modesof the reconfigurable antenna and is selected to achieve at least athreshold level of passive suppression.
 5. The transceiver of claim 1,wherein the antenna controller is further configured to calculate aperformance metric associated with one of the plurality ofreconfigurable antenna modes of the reconfigurable antenna for eachtraining symbol in the set of training symbols.
 6. The transceiver ofclaim 5, wherein, after the training interval, the antenna controller isfurther configured to select a mode of the reconfigurable antenna thatmaximizes or minimizes the performance metric.
 7. The transceiver ofclaim 1, wherein: the first antenna comprises a first reconfigurableantenna having a first plurality of reconfigurable antenna modes; andthe second antenna comprises a second reconfigurable antenna having asecond plurality of reconfigurable antenna modes.
 8. The transceiver ofclaim 7, wherein the antenna controller is further configured to selecta respective mode of the first reconfigurable antenna and select arespective mode of the second reconfigurable antenna for individualtraining symbols in the set of training symbols during the traininginterval.
 9. The transceiver of claim 1, wherein the transmit chain isconfigured to transmit and the receive chain is configured to receiveover a same carrier frequency simultaneously.
 10. A method ofreconfiguring a transceiver, comprising: transmitting, with a transmitchain of the transceiver, a set of training symbols during a traininginterval; selecting, with an antenna controller, a respective mode of areconfigurable antenna of the transceiver for individual trainingsymbols in the set of training symbols, the reconfigurable antennahaving a plurality of reconfigurable antenna modes; calculating, withthe antenna controller, a performance metric associated with one of theplurality of reconfigurable antenna modes of the reconfigurable antennafor each of the set of training symbols; and based on the performancemetric, selecting, with the antenna controller, a mode of thereconfigurable antenna for use during a data transmission interval. 11.The method of claim 10, further comprising selecting the mode of thereconfigurable antenna for use during the data transmission intervalbased further upon a threshold received Signal-of-Interest to InterfererRatio (SIR) of the transceiver.
 12. The method of claim 10, wherein anumber of the set of training symbols is equal to a number of theplurality of reconfigurable antenna modes of the reconfigurable antenna.13. The method of claim 10, wherein a number of the set of trainingsymbols is less than a number of the plurality of reconfigurable antennamodes of the reconfigurable antenna and is selected to achieve at leasta threshold level of passive suppression.
 14. The method of claim 10,further comprising: receiving, with a receive chain of the transceiver,the set of training symbols during the training interval; selecting,with the antenna controller, a respective mode of a secondreconfigurable antenna of the transceiver for individual trainingsymbols in the set of training symbols.
 15. A transceiver, comprising: adigital signal processor configured to transmit a set of trainingsymbols during a training interval; a transmit chain configured totransmit the set of training symbols over an antenna, the antenna havinga plurality of reconfigurable antenna modes; a receive chain configuredto receive at least one of the training symbols as self-interference;and an antenna controller configured to: select a respective mode of thereconfigurable antenna for individual training symbols in the set oftraining symbols during the training interval to calculate a performancemetric; and set a mode of the reconfigurable antenna to reduce theself-interference based on the performance metric during a datatransmission interval.
 16. The transceiver of claim 15, wherein a numberof the set of training symbols is less than a number of the plurality ofreconfigurable antenna modes of the antenna reconfigurable antenna andis selected to achieve at least a threshold level of passivesuppression.
 17. The transceiver of claim 15, wherein the antennacontroller is further configured to calculate a performance metricassociated with one of the plurality of reconfigurable antenna modes ofthe reconfigurable antenna for each training symbol in the set oftraining symbols.
 18. The transceiver of claim 15, wherein, after thetraining interval, the antenna controller is further configured toselect a mode of the reconfigurable antenna that maximizes or minimizesthe performance metric.
 19. The transceiver of claim 15, wherein thereceive chain is configured to receive at least one of the trainingsymbols over a second antenna, the second antenna having a secondplurality of reconfigurable antenna modes.
 20. The transceiver of claim7, wherein the antenna controller is further configured to select arespective mode of the first antenna and select a respective mode of thesecond antenna for individual training symbols in the set of trainingsymbols during the training interval.